Hello.
Welcome to the 2022 Power Supply Design Seminar.
The presentation topic is isolated gate-driver bias supply
design considerations.
My name is Brent McDonald.
I'm a system engineer inside our Power Supply Design Services
Group.
Here is the agenda for the presentation.
First I'll give a brief introduction
of the inverter and isolated gate-driver bias supply
architectures.
Followed by that, we'll discuss several different ways
of creating isolated bias supplies, including the control
methods, different topologies, as well as the transformer
construction methods.
Based on the comparison of different isolated bias supply
solutions, the LLC-based open-loop isolated bias supply
is going to be introduced.
The discussion covers the circuit operation principles
and the circuit variations, its voltage regulation is derived,
and the methods of creating multiple outputs are discussed.
The design procedure is also introduced.
Finally, we'll give some performance demonstration
of the LLC-based open-loop isolated bias supply.
The isolated gate driver is a key component
in many power electronic systems, including the traction
inverter in the electric vehicle--
the motor drive, the UPS system, as well as
in the onboard charger of an electric car.
As shown in this picture, the isolated gate drivers
are normally located very close to the inverter power
stage and allow the MCU to control the power stage.
Here shows an example of the automotive traction inverter,
how the circuits are biased and controlled.
There are two power sources in this system--
the low-voltage battery and the high-voltage battery.
The inverter is directly powered by the high-voltage battery,
while the control circuit is biased
by both the high-voltage and low-voltage batteries.
The high-voltage is processed by an isolated DC-to-DC converter,
and then [? org ?] together with the low-voltage battery
to create a reliable bus voltage and then processed
to the other regulated bus voltages.
The control signals coming from the MCU
are passed through the isolated gate driver
to control the inverter power stages.
You can see the three low-side drivers share the same ground,
but each of the high-side drivers has its own ground.
Given that there are so many grounds in the system,
there is a need to create many isolated bias supplies
for the isolated gate drivers.
How do we do that?
To design the isolated bias supplies,
first we need to understand the requirements, including
output voltages, voltage regulation
requirements, and power levels.
In many designs, the gate drivers
often have positive and negative supplies,
as shown in this picture.
The gate-driver loss can be calculated
using the driving voltage, the gate charge, and the switching
frequency.
The driving voltage requirements are
different for different power devices.
Silicon carbide and silicon MOSFETs are shown as an example.
The plot shows the driver voltage impact
on the RDS(on) of these devices.
The curves show the normalized RDS(on)
for four different MOSFETs, with their RDS(on) measured
at 20 volts gate voltage.
With the increase of their gate voltage, the RDS(on) reduces.
It can be noticed that for the silicon
FET, when the gate voltage is above 10 volts,
the RDS(on) saturates.
Further increases in the gate voltage
will not reduce the conduction loss and the gate-driver loss
increases.
This means the driving voltage for the silicon FET
should only be slightly higher than 10 volts
to get the lowest conduction loss and driver loss.
Normally, we see people drive these devices
with 12 or 15 volts of gate voltage.
And there is not much need for the gate voltage
to be very accurately controlled.
For the silicon carbide MOSFETs, the behavior is quite different.
Their RDS(on) keeps reducing with the increasing of the gate
voltage.
There is no sign of saturation.
This means that the dry voltage needs to be as high as possible
to reduce the conduction loss.
This driver voltage could be very close to the gate breakdown
voltage.
Therefore, you can see the gate-driver voltage
is 18 or 20 volts with a tight regulation requirement,
like somewhere between 1% to 5%.
The IGBT behavior is similar to a silicon FET.
It is normally driven by a 15 or 18 volt gate voltage,
with less concerns on the gate voltage accuracy.
The negative voltage is often needed
to speed up the turn-off process and reduce the switching loss.
Or, it is also added to prevent shoot-through.
This table summarizes the typical gate-driver voltage
requirements for different devices.
After the power requirement, the bias supply architecture
needs to be determined.
There are six isolated gate drivers
in a three-phase inverter.
The centralized architecture creates all the isolated bias
supply rails using a single converter.
This gives a lowest cost solution
because only one power supply is needed.
However, the transformer needs to handle the total power
of all the gate drivers.
The transformer can be heavy and bulky.
It becomes a challenge when the system needs
to pass the vibration test.
Instead of using a single converter,
several converters can be used, all three low-side devices
sharing the same ground.
They can be powered from a single power supply.
Each of the three high-side drivers has its own ground.
Each one of them needs its own isolated bias supply.
We'll call this a semi-distributed architecture.
You can go one step further and make a fully distributed
architecture.
Each isolated gate driver is biased by its own bias supply.
Comparing these three architectures,
we can see that the centralized system has the lowest cost.
But the bulky and heavy transformer
makes it more difficult to pass the vibration test.
Another concern is the fault management.
Because everything comes from a single power supply,
once it fails, everything fails.
The distributed system goes to another extreme.
It has sixth power supplies and it carries more cost.
However, the transformer is spread out to six transformers.
And it is much easier to pass the vibration test.
Besides, it has six individual power supplies.
One power supply failure won't cause the entire system failure.
The fault management becomes much easier.
The semi-distributed system gives
a trade-off between the cost and reliability.
It becomes a popular solution for the automotive industry.
After selecting the appropriate system architecture,
the next question needs to be answered
is, how to control the output voltage.
For an isolated power supply, the most popular control method
is to use the closed loop control
with secondary-side feedback.
The output voltage is directly sensed on the secondary side,
passed through the control loop, then the control signal
is transferred through the optocoupler to the primary side.
And the primary side controller controls the duty cycle
or switching frequency to regulate the output voltage.
This control method directly senses the output voltage
with closed loop control.
It gives a well-regulated output.
Its input can have a wide range.
No pre-regulator is needed.
However, it requires a lot of components for the sensing,
control loop, and isolation.
Besides, the optocoupler generally
has a lower reliability.
People try to avoid using it in the applications
where reliability is critical--
for example, in an automotive system.
To improve the reliability, the optocoupler
can be removed by using the primary-side feedback method.
Instead of directly sensing the output voltage,
the output voltage is sensed on the primary side
through transformer coupling.
The output voltage can then be regulated on the primary side
without using an optocoupler.
Since it can still sense the output voltage,
it can take a wide input voltage range
without requiring a pre-regulator.
Because the output voltage is sensed indirectly,
the voltage regulation performance
is not as good as the secondary-side control.
But it could be good enough for most
of the gate-driver application.
To improve the output voltage sensing accuracy,
the converter has to operate in discontinuous conduction mode
and the output voltage has to be sensed at a specific point.
This makes the voltage sampling method
more susceptible to noise.
Another way of controlling the output
is to use an open-loop control for certain type of power
converters when it's operation condition is fixed.
For example, 50% duty cycle or fixed switching frequency,
there is a fixed relationship between the input
voltage and the output voltage.
And the power supply can be controlled with open loop.
With fixed input voltage, a fixed output voltage
can be created.
Apparently, this power supply needs a fixed input voltage.
With a variable input voltage, a pre-regulator is needed.
Also, the output voltage is unregulated.
It changes slightly with the load condition
depending on how much circuit loss it has.
However, because the control method
doesn't need to sense the output voltage
or have the control loop, it is more immune to noise
and always stable.
It also uses less components.
The open-loop control provides a low-cost and reliable solution
for the isolated gate-driver bias supply.
And it is more attractive in the noisy environment,
such as the industrial or automotive application.
The next step is to choose the right topology.
The isolated bias supply is basically
an isolated DC-to-DC converter.
Many topologies can be used.
Here, we will give several examples.
The flyback converter is the simplest isolated DC-to-DC
converter.
It uses one switch, one diode, and one transformer
to create the isolated bias supply.
It can easily create multiple outputs
by adding more windings to the transformer.
Each output voltage is proportional to the turns ratio.
And it is very suitable for the centralized architecture.
Traditionally, multiple output flyback converters
have a concern about cross regulation.
When the load conditions are different,
the output voltage regulation could be bad.
However, this is less of an issue
in the isolated gate-driver application
because the gate-driver loads are normally well-balanced.
The flyback converter can be controlled
with secondary-side or primary-side feedback.
Normally, we are not using flyback converters
with open open-loop control.
The push-pull converter is another popular choice
because of its simplicity.
It uses a bit more components compared
with the flyback converter.
However, the transformer utilization is better.
It can be used with open-loop control.
When both switches operate with 50% complementary duty cycle,
the output voltage is connected with the input voltage
directly through the transformer.
Therefore, the relationship between the input and output
voltage is simply the transformer turns ratio.
It can also be used with closed-loop control.
In this case, the duty cycle will be less than 50%
to regulate the output voltage and an output filter inductor
is required.
Another option is to use a half-bridge or full-bridge
converter.
Here I show a half-bridge converter.
It is very similar to the push-pull converter
with 50% duty cycle operation.
The output voltage is connected with the input voltage
through the transformer.
There is a fixed relationship between the input and output
voltage.
This circuit can be implemented with a simple gate-driver IC
with a timer to create 50% duty cycle.
Both the push-pull and half or full-bridge converters
are good for the distributed and semi-distributed architectures.
Regardless of which circuit topology we choose,
the transformer is a key element in the isolated bias supply.
The transformer parameters can largely
impact the overall system EMI performance.
Here, we use a three-phase inverter as an example.
Looking at the high-side gate driver of the first leg,
the isolated bias supply gets its power
from the low-voltage supply and transfers it to the high side.
Its output voltage ground is connected
with the inverter switch node.
Looking into the details of this isolated bias supply, when
the output ground fluctuates with the inverter power stage
switch node, the high dv/dt can be
coupled through the parasitic capacitor
between the transformer primary and secondary sides.
We often call this cap an interwinding capacitor.
This current can cause high EMI noise, extra power loss.
And the noise current can potentially
cause a CMTI issue, also known as Common-Mode Transient
Immunity issue.
The controller could also malfunction or trigger
protection mechanisms because of this noise current.
The noise becomes more severe when the wide bandgap
devices are used, such as silicon carbide or GaN.
Because the dv/dt is much higher,
we often hear of 50 volts per nanosecond
or even 100 volts per nanosecond of voltage slew rate
with these devices.
Here is some simple math.
With 100 volts per nanoseconds of dv/dt,
even a 1 picofarad parasitic cap can cause
0.1 amps of noise current.
And we normally see about 20 to 30 picofarads
of interwinding capacitance for a flyback transformer.
It is clear that a good, isolated bias
supply for the isolated gate driver
should have a very low interwinding capacitance.
So how do we make the interwinding capacitance
smaller?
Here we see a typical way of making a transformer.
Outside is the core and inside are the bobbin and windings.
The windings are put on the bobbin by layers--
first, the primary layer, then the insulator,
then the secondary layer.
The capacitance can be estimated by this equation.
Reducing the surface area or increasing the distance
can reduce the capacitance.
One easy way is to increase the insulator thickness--
basically, increase the distance.
This is less effective because of the large surface area.
Another way of reducing the capacitance
is to use a split-chamber bobbin, as shown here.
The primary side winding and secondary side windings
are put in separate chambers.
They are physically separated apart.
The surface area is reduced and the distance is increased.
It can achieve a much smaller interwinding capacitance.
Regardless of which method is used,
increases in the winding distance
reduces the capacitance, but it also
reduces the coupling between the primary side and secondary side
windings, which means the leakage inductance will also
be increased.
This slide summarizes different transformers
with different construction methods
for different topologies.
With respect to the interwinding capacitance,
the flyback, the half-bridge, and the full-bridge transformers
have relatively large values, around 20 picofarads.
The push-pull transformer uses a toroid core
and has an interwinding capacitance
around 10 picofarads.
When the split-chamber bobbin is used
the interwinding capacitance can be reduced
to be less than 2 picofarads.
This is an order of magnitude amplitude reduction compared
with the flyback transformer.
The lower interwinding capacitance
means the design can easily pass the high dv/dt CMTI test.
It also helps achieve noise immunity of higher
than 150 volts per nanoseconds.
This is better than the other transformers.
Not only the reduction of parasitic capacitance,
the split-chamber bobbin also allows a fully automated
manufacturing process.
This allows the transformer cost to be 15% to 30% lower compared
with the other transformers.
As we discussed earlier, the reduction
of the interwinding capacitance has the penalty of increasing
leakage inductance.
The split-chamber bobbin transformer
has a much higher leakage inductance.
The push-pull transformer uses bifilar
to achieve a very low leakage inductance.
In terms of regulation, the PSR flyback
allows the closed-loop control to achieve the best regulation
performance.
The three-winding flyback uses the rectified aux winding
voltage as the feedback.
It gets good regulation performance,
but not as good as a true PSR flyback controller.
The other transformers are used for open-loop control.
The voltage regulation performance is not as good.
To get better EMI performance, we
need the buyer supply to have a minimum interwinding
capacitance.
As a result, the transformer will have a much larger leakage
inductance.
However, not every topology is able to use a transformer
with large leakage inductance.
For example, in flyback or push-pull converters,
the transformer is in series with the switch.
Once the switch turns off, the energy
stored in the leakage inductance has no place to go.
It can't be transferred to the secondary side.
The leakage inductor resonates with the switch node capacitor,
causing more EMI noise, more loss, and more voltage stress
on the devices because of the voltage ringing.
Furthermore, a snubber is often added
to manage the leakage energy.
This could also increase the solution cost.
Therefore, for these topologies, the leakage has to be minimized,
which means the transformer interwinding
capacitances can't be reduced.
In a half-bridge converter, the transformer
is connected with a half bridge.
The leakage energy can be fully recovered
and the leakage inductor won't cause extra ringing.
However, the leakage inductance still
needs to be minimized for the voltage regulation.
Let's take a look at the half bridge.
The half-bridge converter can be used as a PWM converter
or a resonant converter.
When it is used as a PWM converter,
the leakage inductance is between the input voltage
and the output voltage.
If we draw its equivalent circuit,
basically the leakage inductance is in between the input
and the load.
For different loads, the larger leakage inductance equivalently
gives more impedance to the source, which causes worse load
regulation performance.
In this case, the leakage inductance
still needs to be minimized.
However, when the half-bridge operates
as a resonant converter, we put a resonant cap in series
with the leakage inductance.
When the switching frequency is at the resonant frequency,
the resonant tank impedance is zero.
It can achieve a much better load regulation performance.
Because the resonant capacitor can tune out
the leakage inductance impedance,
transformers used in the resonant half bridge
can be low capacitance and high leakage.
One popular resonant half-bridge converter is the LLC converter.
It gets its name from its resonant components--
L, L, and C.
The two resident inductors can be part of the transformer.
LR is the leakage inductance and LM is the transformer
magnetizing inductance.
The resonant frequency is determined
by the resonant capacitor, CR, and the resonant inductor, LR.
The half bridge operates with a 50% duty cycle.
When it's switching frequency is at the resonant frequency,
the transformer primary side and secondary side currents
are both sinusoidal.
There is a slight phase shift between the primary side current
and secondary site current.
This is caused by the magnetizing current.
The secondary side current is then rectified and filtered out
to create the output voltage.
The voltage gain of this converter
can be defined by this equation.
Here, 2 comes from the half-bridge operation.
The resonant frequency comes from the series,
L and C. The load resistor is converted based
on the sinusoidal current.
And the quality factor is defined
by the typical characteristic impedance
divided by the equivalent load.
The normalized gain curve of the LLC converter is shown here.
The switching frequency is normalized
with the resonant frequency.
And each curve represents different load conditions.
Generally, the voltage gain becomes
higher when the switching frequency becomes higher.
However, noticing this point, right at the resonant frequency,
the gain is equal to 1, regardless of load conditions.
This is because at the resonant frequency
the impedance of the resonant tank is equal to zero.
The input and output voltages are shorted
through the transformer.
This property allows us to use the converter with a fixed
frequency open-loop control.
With low capacitance, large leakage transformer,
the leakage inductor can be conveniently used
as the resonant inductor.
Looking at this set of curves, gain
is only the same when the switching frequency
is equal to the resonant frequency.
In reality, it is impossible to always achieve that.
What if the switching frequency is not
equal to the resonant frequency?
Can we still use the topology?
First, let's take a look at the LLC gain curves
with different leakage indulgences.
Here, we define a new parameter, LN.
LN is the ratio between magnetizing inductance
and the leakage inductance.
Or, another way to put it, it's 1
over the percentage of leakage.
Here, we show two sets of curves,
one with a relatively small LN equal to 1.
You can see that the gain curve is
very sensitive to the switching frequency.
A small frequency change could result in large voltage gain
changes.
However, when the LN is relatively large,
say equal to 15, the gain curve becomes much flatter.
This means that the voltage gain is
less sensitive to the frequency change.
The circuit can tolerate frequency error
from the component tolerances.
Or, in other words, when the magnetizing inductor
is much larger than the resonant inductor,
around the resonant frequency the gain curve is flat.
We should use the LLC converter with such a transformer.
Other than circuit parameters, we
can also adjust the circuit configuration
to make the voltage gain less sensitive to the frequency
shift.
Here are shown two circuit schematics.
They are almost identical.
There is a slight difference between the two, the location
of the resonant capacitor.
The circuit on the left has its resonant capacitor
located on the primary side, while the circuit on the right
has its resonant capacitor located on the secondary side.
When we do the gain analysis of these two converters,
assuming a 24-volt input and a one-to-one transformer turns
ratio for the primary-side resonant,
the voltage gain changes with frequency.
While for the secondary-side resonant,
the voltage gain is almost flat when the switching frequency
is below the resonant frequency, which
means the secondary-side operation is even less sensitive
to the frequency tolerances.
But why?
Comparing the operation waveforms,
these two circuits have very similar switch node waveforms
and current waveforms.
When the switching frequency is below the resonant frequency,
however, the difference happens during this period
when the resonance stops.
For the primary-side resonant circuit, during this period,
the secondary-side circuit is off.
The magnetizing inductor will resonate
with the resonant capacitor and transfer its energy
to the resonant capacitor.
In the next half-switching cycle,
the energy is transferred to the secondary side.
This causes the gain to become higher.
For the secondary-side resonant circuit,
at the same period, because the resonant cap is located
on the secondary side, the energy
stored in the magnetizing inductor
is fed back to the source, but not to the secondary side.
This means the voltage gain won't go up.
Once the operation principle is clear,
we can see that the secondary-side resonant circuit
is less sensitive to the switching frequency errors.
There are a few variations of the half-bridge LLC converter--
through the transformer, the rectification method, as well as
the resonant capacitor location.
Regardless which configuration we choose,
the filter cap should always be much larger
than the resonant cap to prevent its interaction
with the resonance.
The transformer can use two simple windings
or a center-tap structure.
Since the LLC converter uses a capacitive filter
at its output, the same as a flyback converter,
it can also be used with multiple secondary-side windings
to create multiple output voltages.
The rectification method, we can use the full-bridge rectifier
as shown in the original circuit or use
the center tap with the center-tap transformer.
We can use the voltage doubler structure
with one or two capacitors.
The one capacitor is basically those two capacitors
in parallel.
At low power levels, there is really not much difference
between these two structures.
You can pick up one based on the size and cost considerations.
The resonant cap location has been discussed
in the previous slide.
Considering all these options, the two-winding transformer,
voltage doubler, and the secondary-side resonant
could be a preferred solution.
The voltage doubler also brings an additional benefit.
Because of the half bridge, the transformer primary only
sees half of the input voltage.
When the voltage doubler is used,
the transformer secondary side only
sees half of the output voltage.
This means the voltage conversion ratio is simply
the transformer turns ratio.
This makes the math easier.
The voltage gain analysis done so far
is based on the ideal conditions.
The open-loop LLC voltage regulation
can be calculated considering all the lossy elements
in the circuit.
Here, I show a half-bridge LLC resonant converter
with a voltage doubler output stage.
When we consider the losses, the circuit
can be redrawn like this.
The switch is replaced by an ideal switch
with its on-state resistance.
RP represents the transformer primary-side resistance.
And RS represents the transformer
secondary-side resistance and other resistive conduction
losses from the capacitor and diode.
Because the primary side is a half bridge,
we can represent the input voltage as a square wave.
Same thing for the secondary side--
represent the secondary side using a square wave.
At the same time, we can reflect everything from the primary
to the secondary side to simplify the math.
The square wave is still quite difficult to analyze,
given the circuit operates very close to the resonant frequency
and the resonant tank impedance increases rapidly
when the frequency moves away from the resonant frequency.
Basically, only the fundamental harmonic of the square wave
can be transferred through the circuit.
We can replace the square wave with its fundamental frequency
component and represent the output with an equivalent
resistor, and the circuit becomes this.
Now the circuit can be analyzed easily
and the output voltage can be written in this equation.
It has two terms--
a DC voltage and an impedance.
So the output behaves as a DC source
with finite internal impedance.
The DC voltage is the input voltage
reflected to the secondary side, then minus the diode drop.
The impedance represents the circuit loss.
It's the combination of the switch on-state resistance,
the AC resistance from the transformer,
and the resistive loss from the capacitor and the diode.
When using the closed-loop control,
the internal impedance can be tuned out.
When using the open-loop control,
we have to live with this impedance.
We can try to minimize the resistive loss
to get lower output impedance and better load regulation.
Even though the open-loop LLC output
can see some load regulation, the load regulation
behaves differently for different load conditions.
This plot shows a curve that represents a typical output
voltage at different load conditions.
When the load increases, the output voltage drops.
When we make a large number of these circuits
due to component tolerances, the output voltage
will be within these bands.
Some will be higher and some will be lower.
As we discussed earlier, the gate-drive power
is determined by the driver voltage, gate charge,
and the switching frequency.
Given the driver voltage and gate charges fixed,
the load is only determined by the switching frequency.
For a fixed switching frequency inverter,
the gate-drive power is fixed and the output voltage
varies from here to here.
It is much smaller compared with the entire load
range and tolerances.
The voltage regulation can be very tight.
During standby mode, the gate-drive losses
become a minimum.
The output voltage tends to go higher
due to the junction capacitance of the diode in the bias supply.
An additional dummy load or Zener clamp
might be needed to prevent that voltage from becoming too high.
For a variable frequency inverter,
the gate-drive load varies and the output voltage varies more.
The best way to analyze the output voltage regulation,
considering the component tolerances,
would be to use a Monte Carlo simulation.
This circuit is used to simulate the output voltage tolerances.
Here are some assumptions.
We'll use a 24-volt input with the transformer turns ratio
assumed to be one-to-one.
The leakage inductance is distributed
evenly on the primary side and secondary side
with a 20% tolerance.
The tolerance of the magnetizing inductor
is 40%, 6% switching frequency tolerance,
and 10% tolerance for the resonant capacitor.
With all these tolerances, for each load condition,
we can see the output voltage varies about 0.2 volts
at a 22-volt output.
This shows the output voltage is really not
that sensitive to the circuit parameter shifts.
Next, let's take a look at how to design this open-loop LLC
converter.
Traditionally, the LLC converter's design
is complicated and challenging.
This is because people try to optimize
the efficiency, the power density, the input and output
voltage range, et cetera.
It is quite different in the isolated bias supply.
The bias supply is a small portion of a large system.
The efficiency is not critical because the loss is negligible
compared with the overall system losses.
Optimizing the efficiency can't improve the overall system
performance.
The size of the converter is normally
dominated by the isolation requirement.
The size reduction is mainly achieved
through the minimization of the number of components
instead of reducing the component size.
Furthermore, because of the open-loop control,
there is no need to consider voltage gain.
The design of this LLC converter can be much simpler.
First, based on the input voltage and output voltage
requirement, the transformer turns
ratio can be determined by considering
the extra headroom needed.
Once the transformer turns ratio is determined, with the load
current requirement, the transformer primary
and secondary-side RMS currents can be calculated.
The input voltage, together with the switching frequency,
can determine the volt second rating of the transformer.
Using a split-chamber bobbin without an air gap,
the transformer can be designed to achieve the desired
magnetizing inductance.
In this equation, the magnetizing inductance design
target is to achieve soft switching.
However, for this low-voltage input converter,
soft switching is really not that critical.
Even without ZVS, the switching loss is still small.
This value gives a design target to start with.
We are not giving a leakage inductance design target.
Instead, once the transformer is designed,
we measure the leakage inductance,
then match the leakage inductance
with a resonant capacitor, set up the resonant frequency
at about 10% above the switching frequency.
The open-loop LLC converter can give a single output voltage
for the isolated gate drivers.
Often positive and negative rails are needed.
We can use some simple circuits to split the single output
voltage into positive and negative rails.
The Zener split is a simple solution.
It uses a Zener diode to create the negative voltage.
And the rest of the voltage becomes the positive rail.
Clearly, this gives a low-cost solution.
However, both the positive and negative rails are unregulated.
The Zener diode can also be put on top
to create the positive rail and let the remaining voltage become
the negative rail.
To improve the accuracy, a shunt regulator
can be used to replace the Zener.
This requires a higher cost, but the negative rail
becomes regulated.
The positive rail is still unregulated.
One step further, we can add a linear regulator
on the positive rail.
This allows us to have a regulated positive
rail and negative rail.
But it requires extra cost and less efficiency.
Based on all these analyses, the UCC25800-Q1 open-loop LLC
transformer driver was developed.
It integrates a half-bridge power stage with a 1-amp peak
current capability, a frequency setting resistor to set
the switching frequency between 100 kilohertz to 1 megahertz.
The switching frequency becomes 1.2 megahertz
if the pin is left open.
It also has a built-in soft start of 1.5 milliseconds.
The overcurrent protection and maximum dead time
can be set up through this pin.
The overcurrent level can be programmed for different power
levels with overload and short-circuit protections.
The maximum dead time is programmed through the divider
ratio.
The IC can achieve automatic dead time adjustment
within the programmed maximum dead time
to achieve the best ZVS performance.
The IC also has a disable pin, allowing
it to be disabled through an external signal.
This pin also serves as a fault code output pin.
It sends out a series of pulses to let
the system know what protection feature is triggered.
You can read the pulse number to help
identify the cause of the protection
and simplify the debugging process.
Other than the program switching frequency,
it can also be synchronized with an external clock signal.
A special handoff mechanism was designed
to ensure smooth transition.
Here shows the transition and the circuit operation
is not disturbed during the frequency handoff.
It also has a full set of protection features integrated
inside, including input overvoltage
protection, thermal shutdown, and pin open-short protections.
Here is some measure data.
The measurements are based on the UCC25800 open-loop LLC EVM
with the LM5156 SEPIC pre-regulator.
The SEPIC circuit takes a 6 to 26-volt input voltage,
converts it to a regulated 15-volt intermediate bus.
Then the open-loop LLC converts the 15, plus 18 volts
and minus 5 volts, with 85 milliamps of output current.
The SEPIC switching frequency is at 2.2 megahertz
and the LLC switching frequency is 500 kilohertz.
A post-regulator circuit using a shunt regulator
and a linear regulator is implemented on this board.
Before, the post-regulator, the total output voltage
varies between 24 volts and 25 volts.
With the post-regulator, 1% load regulation can be achieved.
With soft switching sinusoidal current and low
interwinding capacitance, the circuit
can easily pass the CISPR 25, Class 5 EMI standard.
The open-loop LLC can also be used
for creating multiple outputs for centralized architectures.
There are two ways of creating multiple outputs--
using a transformer with multiple secondary-side windings
or using a half bridge to drive multiple transformers.
For the multiple secondary-side windings,
only one transformer is needed.
However, due to the complex coupling on the secondary side,
primary resonant is preferred.
In this configuration, even though we still
reduce the coupling between the primary side
and the secondary side.
The secondary-side windings need to be coupled well.
This still has the noise coupling
among different outputs.
Or, we can use one half bridge to drive multiple transformers.
In this case, the half bridge only creates a square wave.
And each LLC operates independently.
The secondary-side resonant can be used.
And the coupling, among all the outputs, is minimized.
This is an example of using multiple transformer.
The primary side is using the UCC25800 power stage.
It drives three transformers and secondary-side circuits.
The three output voltages are measured with 25 volts input.
Three well-matched output voltages are created.
The LLC converter has soft switching,
lower dv/dt on the switch node, sinusoidally shaped current,
and minimum interwinding capacitance.
We are expecting it to have a better EMI performance
all on its own, not to mention a much smaller noise coupling
from the inverter power stage.
Here shows the EMI performance comparisons among 5-volt input,
push-pull, 24-volt input flyback, a 24-volt input LLC,
without adding the EMI filter, just to see the bare noise.
It can be seen that the LLC converter has a much
lower high frequency EMI noise.
Normally, the low frequency EMI can be handled easily
through the EMI filter.
But the high frequency EMI noise is much more
difficult to manage.
The LLC converter provides a much better EMI noise
performance.
We also put the circuit into a CMTI test.
CMTI stands for Common Mode Transient Immunity.
As we discussed earlier, when the bias supply is
used to power the high-side gate driver,
the high dv/dt from the inverter power stage
can be coupled through the transformer interwinding
capacitor to cause common-mode current
and cause different issues.
This dv/dt can be well above 50 volts per nanosecond.
This condition can be simulated by adding
a pulse voltage between the bias supply input and the output
grounds.
We call this voltage a strike voltage.
This slide shows the CMTI performance.
The blue waveform is the LLC switch node voltage.
And the yellow curve is the strike voltage.
Looking into the details, we can see
the strike voltage has a slew rate of 165 volts per nanosecond
at its rising edge, and 155 volts per nanosecond slew
rate at its falling edge.
The strike voltage is from negative 500 to plus-500 volts.
Looking at the LLC switch node behavior, right
around the strike edge we can see with the higher
than 150 volt per nanosecond voltage slew rate,
the LLC converter operation is not disturbed.
The circuit can withstand this high dv/dt easily.
Other than the EVM, there are some reference
designs available to assist your design.
The PMP22835 is a 24 volt in, 18 volt, and minus-5 volt output,
6 watt, 1 megahertz design.
The PMP22930 takes 15 volts in, creates
a plus-15 and minus-4 volt, 2.6 watt output with a 600 kilohertz
switching frequency.
The PMP23061 has a boost regulator that regulates a 6
to 28-volt input voltage to a 30-volt intermediate bus
voltage.
One UCC25800 then converts it to four 18-volt outputs,
one 3-watt output, and three 1-watt outputs.
It has 500 kilohertz switching frequency.
This is designed to power up three-phase inverter gate
drivers.
Here's a summary of what we discussed in this presentation.
An isolated bias supply is necessary for the biasing
of isolated gate drivers in inverters.
Open-loop control provides a reliable solution
and is less noise sensitive.
The LLC topology is able to use the transformer with large
leakage inductance and minimize the transformer's
primary-to-secondary-side parasitic capacitances.
It minimizes the noise coupling.
The open-loop LLC converter provides
a simple, robust solution with less EMI, high CMTI,
good voltage regulation, and multiple output capability.
I want to thank you for your time and attention.